Middle-East Journal of Scientific Research 20 (12): 2252-2257, 2014 ISSN 1990-9233 © IDOSI Publications, 2014 DOI: 10.5829/idosi.mejsr.2014.20.12.315 A Novel Low Power CMOS PLL for Clock Generation A. Geetha Department of ETE, Bharath Univeristy, India Abstract: In this paper we propose a Low Power Phase Locked Loop (PLL) using transmission gate logic (TG). The proposed PLL has a simpler structure using very less number of transistors compared to the conventional PLL and results in better performance than any other published PLL’s due to no DC leakage currents and reduced transistor count. The PLL has a dead zone less than 0.01ns in the phase characteristics and has low phase sensitivity errors. The PLL is independent from the duty cycle of input signals and the effects of clock feed – through are reduced by separating the output stage from UP and DOWN signal. The PLL is designed using 0.8 µm CMOS technology. We compared the proposed PLL with the conventional PLL with respect to performance and power consumption. The Results show 30% reduction in power consumption over a conventional PLL with comparable performance. Key words: Low Power PLL Phase and Frequency Detector INTRODUCTION The minimum channel length of the transistor will be scaled down to 0.065 um in 2007, according to the roadmap of semiconductors. In addition to this downscaling, today’s system-on-chip (SoC) trend forces analog and digital integrated circuits (ICs) to be integrated on a single chip called the complete SoC. At present, there are many demands on the complete SoC in wireless and broadband communications – wireless networking (WLAN, voice/data communication and Bluetooth), wired communication (WAN and LAN) and consumer electronics (DVD, MP3, digital cameras, video games and so on). Therefore, as one of the mixed-signal Ics, Phase-Locked Loop (PLL) must follow this complete SoC trend. This chapter introduces the challenges in designing PLL and possible solid-state technologies for the complete SoC trend. The Low Power phase locked loop (PLL) circuit is used to synchronize an output signal, which is usually generated by an oscillator, with a reference or input signal in Frequency as well as in phase. In the synchronized state, the difference (error) between the reference and the oscillator output is zero or at least very small. So it is called ‘locked’. The whole circuit consists of three main parts. They are phase & frequency detector (PFD), charge Pump, loop filter (LP) voltage control oscillator (VCO) and divider. The diagram of the PLL is shown in Figure 1. Mixed signal circuits The main motivation for this paper is to design a PLL that will allow on-chip direct digitization of a wideband RF signal. Yet one of the major challenges in developing the complete SoC product for the wireless digital network market is the integration of radio frequency (RF) analog circuit devices, which are mostly passive discrete devices. Eventually, we want to replace the passive devices with active devices. We also want to replace analog designs with full-digital implementations. To do this, a suitable PLLs architecture, concentrating on low jitter and power, for the complete SoC should be devised. In section II, the proposed Novel Low Power Phase Locked Loop is described. The results of proposed PLL circuit are presented and comparisons are made in section III. In section IV, draws a conclusion. Design of Novel Low Power PLL: This section provides detailed information about the proposed PLL, including its major components. Also important characteristics of the PLL are introduced for simulation and analysis, which will be described in this section. Phase and Frequency Detector: The input phase errors are detected by Phase and Frequency Detector (PFD). These phase or frequency errors are converted into current or voltage to control the output frequency of Voltage Controlled Oscillator (VCO) by charge pump in a charge pump PLL. PFD detects a phase error between the Corresponding Author: A. Geetha, Department of ETE, Bharath Univeristy, India. 2252 Middle-East J. Sci. Res., 20 (12): 2252-2257, 2014 Fig. 1: Low Power PLL Fig. 2: Conventional PFD Schematic reference signal and the output signal of PLL and the error detection range can be extended with PFD. A conventional CMOS PFD [1-4] is shown in Fig. 2 has large dead zone in phase characteristics at the steady state which generates a large jitter in clocked state in PLL and consumes large amount of power, which cannot be avoided in high frequency operations because internal nodes of PFD are not completely pull up or pull down. The additional prescaler circuits can be added to lower the frequency of the input signals. However, as the division ratio increases, the steady state phase error will increase. The capture range of PLL is determined by the error detection range of PFD. A conventional CMOS PFD has no limit to the error detection range. Therefore, the capture range of PLL is only limited by the Voltage Controlled Oscillator (VCO) output frequency range . Our novel PFD uses a two multiplexers, as the phase difference between the input decreases, the pulse width on UP or DOWN also decreases. These simultaneous up and down signals in the steady state of the PLL create a short circuit in the charge pump which results in a perturbation on LF voltage and produce jitter. To limit the LF voltage perturbation without having a dead zone, we have reduced the minimum UP and DOWN pulse widths by reducing the reset delay in the PFD.Moreover, the linearity of the PFD is affected by this minimum pulse width on UP and DOWN when there is no phase difference between the inputs of the PFD.The new PFD has no dead zone and the nonlinearity near the steady state of the PLL is reduced. This implementation also reduces the transistor count to a large extent. The conventional phase and frequency detector (PFD) uses a two D-flip-flop (DFF) structure shown Fig. (1). The DFF has been simplified and implemented using a two 2:1 multiplexer as shown Fig. (4), is optimized to obtain a minimum PFD delay with reduced power consumption. This new PFD schematic has a reduced delay because the number of serial MOS devices in each gate is limited. The new PFD results in a reduced dead zone and non-linearity near the steady state of the PLL. The conventional PFD has the same basic function as the NAND based PFD, but the state assignment is different. Two D flip flops with D=1 are clocked with the clock signals which are compared. If CLK is active before VCO the UP output is generated while DOWN is produced if VCO is active before CLK. As soon as the two clock signals are simultaneously active at the same time, ideally neither up or down has to be set. In such case, the reset signal R has to be fast enough to minimize the UP and DOWN activation time when PLL is in the steady state (perfect synchronization between CLK and VCO). Each DFF used in the conventional PFD uses 19 MOS transistors shown in Fig. 3 Based on this DFF, Fig. 4 shows the PFD according to Fig. 3 depicted uses 42 MOS transistors, the reset signal R is controlled by NQ. In the design of our novel PFD the 2:1 multiplexer replaces the each DFF of Fig. 3, shown in Fig. 5. The PFD using two 2:1 multiplexer uses 9 MOS transistors. The Multiplexers are designed using transmission gate. The design of the new PFD has been performed using an asynchronous race free design method. A detailed description of this design methodology is presented in 2253 Middle-East J. Sci. Res., 20 (12): 2252-2257, 2014 Fig. 3: DFF Using 19 MOS Transistors Fig. 4: Phase And Frequency Detector Fig. 5: Novel PFD Using Multiplexer 2254 Middle-East J. Sci. Res., 20 (12): 2252-2257, 2014  and . A basic schematic of such a circuit is shown in Fig. 5. This circuit has the same basic function as the conventional based PFD.In conventional PFD two D flip flops with D=1 are clocked with the clock signals which are compared. If CLK is active before VCO the UP output is generated while DOWN is produced if VCO is active before CLK. As soon as the two clock signals are simultaneously active at the same time, ideally neither up or down has to be set and a reset signal is given to the DFF and the system is locked as there is perfect synchronization between CLK and VCO [7-11]. The architecture of our novel PFD has one three input NAND gate and two 2:1 multiplexers. One of the input of the NAND gate is always made high and the other input are UP and DOWN signals. The selection input of multiplexer (1) is CLK and of multiplexer (2) is output of VCO. The output of the NAND gate is given to the input i0 of both the multiplexers and the input i1 of the both the multiplexer are grounded. The operation of the our novel PFD, if CLK is active before VCO the UP output is generated because the CLK given to selection input of the Multiplexer (1) is activated before VCO and the UP value is high so the reset value is low which in turn makes the input to be same. while DOWN is produced if VCO is active before CLK because the VCO given to selection input of the Multiplexer (2) is activated before CLK. As soon as the two clock signals are simultaneously active at the same time, ideally neither UP or DOWN has to be set. In such case, the reset signal R is given to the NAND gate becomes high and therefore the output value changes to low which depicts the reset state of DFF. This results in lower jitter and better linearity near steady state without dead zone and reduced power consumption . Charge Pump: A charge pump consists of two switched current sources that pump charge into or out of the loop filter according to two logical inputs. The circuit has three states. Charge pump consists of two PMOS and two NMOS, which are connected serially. Both of the NMOS are in the pull down section and both of the PMOS are in the pull up section. The gate of uppermost PMOS is connected to GND The gate of lowermost NMOS is connected to VDD. The gates of the remaining NMOS and PMOS are connected to the "Down" and "Up" pin of the output of PFD. The width of the uppermost PMOS is 15 um and the length of it is 7.5 um. The width of the other PMOS is 30 um and length of it is 600 nm. The width of the lowermost NMOS is 7.5 um and length of it is also 7.5 um. The width of the other NMOS is 15 um and the length of it is 600 nm. Table 2: Calculated Power At 120 Mhz @1.35V Total number of Transistors Power consumption Conventional PFD Novel PFD using MUX 42 0.324mw 8 0.11mw Table 1: Calculated Area (With Parasitic Coupling Capacitance Extracted From Layout) The Output Peak To Peak Jitter At120mhz @1.35V Conventional PFD Dead Zone (Jitter) Layout Area (mm2) < 10ps 0.21 Novel PFD using MUX < 8ps 0.15 Fig. 6: Charge Pump Low Pass Filter: The Filter used here is a simple low pass Filter. It is comprised of a capacitor and a resistor. The resistor value is 300 k ohm and capacitor is 10 pico farad. The control voltage for the VCO is taken parallel to the capacitor. The values of the component here are taken iteratively. Basically these values allow control voltage to be stable so that small changes or interferences do not affect the locked stage . Voltage Controlled Oscillator: As far as the PSNR and low-supply voltage operation are concerned, the VCO is the most critical block because its internal noise results directly in jitter. Moreover, low-voltage operation limits the design options. To obtain a fully integrated PLL, a current-controlled ring oscillator (CCO) is the basic element of the VCO. It allows low-voltage operation, since no additional capacitances have been used on internal nodes other than the ones created by the inverter devices. To limit the MOS device channel white noise, the current 2255 Middle-East J. Sci. Res., 20 (12): 2252-2257, 2014 Fig. 8: Output of Phase Locked Loop during the locked condition of input and VCO signal. flowing in the CCO must be efficiently high . Allowing a limited amount of white noise in the CCO and given an operating frequency, it can be shown that the voltage across the CCO is minimum and depends only on process parameters which are the best case for a given process. Then as the supply voltage is specified, the maximum saturation voltage of the mirror supplying the current to the CCO is determined. Working down to 1.2V, it is not possible to use a cascoded mirror because the saturation voltage of such a mirror is too high for the expected PSNR. Therefore, to ensure a maximum PSNR even at low-supply voltages, a new circuit called active cascode has been used. The high-frequency PSNR is mainly determined by the ratio of the parasitic capacitance of the mirror MOS devices and the filtering capacitor across the CCO. This last capacitor cannot be increased too much because it introduces a pole in the PLL that can make it unstable. The size of the mirror devices is determined by the maximum allowable saturation voltage. The stability of this active cascode is ensured by the resistor R between the input and output of the mirror. The relation between current and frequency in this CCO is not linear. To a first approximation, the frequency depends on the square root of the current for a voltage across the CCO much larger than the maximum threshold voltage of PMOS or NMOS devices. To obtain a linear gain for the VCO, the voltage to current converter should have a quadratic transfer function. This is achieved by using a MOS device in strong inversion . Divider: Frequency Divider divides the output frequency before feeding it to PFD. It takes input from VCO and divides the frequency, which is the dclock signal. The Frequency Divider designed for the PFD DPLL is programmable. It takes 8 bit input to divide the frequency so the frequency can be divided by 1 to 255 times. The Frequency Divider has three basic parts - an 8-bit synchronous counter, an array of 2input XNOR gates to take input bits and an 8 input Nand gate. The schematic of the Frequency Divider is shown in the Figure 3.6. The Jitter introduced by these dividers is cancelled by a D Flip flop that resynchronizes the output signal of the dividers with the output frequency of the PLL.The divider is implemented using transmission gate, therefore there is drastic reduction in the transistor count. RESULTS AND DISCUSSION Verifications for logic, timing simulations are done by using Modelsim, power estimations and final layout without any DRC and LVS problems using Microwind is also shown in Fig. 9. Fig. 9: Layout Of Low Power CMOS Phase Locked Loop 2256 Middle-East J. Sci. Res., 20 (12): 2252-2257, 2014 3. 4. 5. 6. Fig. 10: Rtl Simulation Of Pll 7. CONCLUSION We proposed a Novel Low Power Phase Locked Loop, the proposed new PLL has simpler structure and lower power dissipation. The dead zone of PLL is less than 0.01ns and error detection range is not limited. With no DC leakage current the power consumption is extenuated and area is drastically reduced. With MOS transistor count reduced are exponentially reduced. The Proposed PLL reduces the delay, which improves the performance of PLL. The designed circuit shows that it is possible to overcome the issue of PLL short settling time by using a very low power PFD operating even in idle mode. So, the recovery time from idle mode to normal mode is virtually zero. The PFD is designed using 0.8 µm CMOS technology. 8. 9. 10. 11. 12. 13. REFERENCES 14. 1. 2. Kaenal, V. and A. Von, 320 Mhz, 1.5 mw @ 1.35V CMOS PLL for microprocessor clock generation in ISSC/C. Montanaro, James, 160 Mhz 0.5W CMOS Risc Processor in ISSCC 96. 15. 2257 Pamel, M. Van, Analysis of charge pump for PLL’s IEEE Trans. July 99. Maxim, A., 2005. A 0.16-2.55-GHz CMOS active clock deskewing PLL using analog phase interpolation, IEEE J. Solid-State Circuits, 40(1): 110-131. Kim, J., J.K. Kim, B.J. Lee, N. Kim, D.K. Jeong and W. Kim., 2006. A 20-GHz phase locked loop for 40-Gb/s serializing rransmitter in 0.13-µm CMOS, IEEE J. Solid-State Circuits, 41: 4. Hung, C.M. and K.O. Kenneth, 2002. “A fully integrated 1.5-V 5.5-GHz CMOS phase locked loop,” IEEE J. Solid-State Circuits, 37(4). Wang, Z., 2005. An analysis of charge-pump phaselocked loops, IEEE Trans. on Circuits and Systems I(52): 10. Hanumolu, P.K., M. Brownlee, K. Mayaram and U.K. Moon, 2004. Analysis of charge pump phaselocked loops, IEEE Trans. on Circuits and Systems I(51): 9. Staszewski, R.B. and P.T. Balsara, 2005. Phasedomain all-digital phase-locked loop, IEEE IEEE Trans. on Circuits and Systems II(52): 3. Razavi, B., 2001. Design of Analog CMOS Integrated Circuits, 2 nd Ed., New York: McGraw-Hill. Chen, W., 2000. The VLSI Handbook. CRC Press, Boca Raton, FL. Baker, R.J., H.W. Li and D.E Boyce, 1998. CMOS Circuit Design, Layout and Simulation. IEEE Press, New York, NY Best, R.E., 1997. Phase-Locked Loops, 3rd Ed. McGraw Hill. Semiconductors, Phillips, 1988. An overview of Phase Locked Loops, AN177, Application Note. Allen, Phillip and Douglas Holberg, 2002. CMOS Analog Circuit Design, Oxford Univ Press.
© Copyright 2018 ExploreDoc